日本音響学会誌
Online ISSN : 2432-2040
Print ISSN : 0369-4232
30 巻, 10 号
選択された号の論文の9件中1~9を表示しています
  • 柴山 乾夫
    原稿種別: 本文
    1974 年 30 巻 10 号 p. 529-530
    発行日: 1974/10/01
    公開日: 2017/06/02
    ジャーナル フリー
  • 杉山 精, 吉川 昭吉郎
    原稿種別: 本文
    1974 年 30 巻 10 号 p. 531-540
    発行日: 1974/10/01
    公開日: 2017/06/02
    ジャーナル フリー
    In the present paper, a unified analysis is given for studying the characteristics of surface wave delay lines using interdigital transducers or phase-coded transducer configurations for dc pulse transmission based on equivalent circuits and Laplace transform calculus. A general idea of the transmission characteristics of a delay line using an interdigital transducer may be obtained by calcuulating the short circuit transfer admittance Y_<TS>. If the number of exciting and detecting electrode pairs is one, the transfer admittance is expressed by Eq. (1). Fig. 2 shows the characteristics of a line in the time domain expressed in terms of Laplace transforms by replacing (jω) with the by operator p in Eq. (1). If the number of exciting and detecting electrode pairs is large, the transfer admittance is obtained from the superposition of the transfer admittance denoted by Eq. (1). For an interdigital exciting transducer composed of M periodic sections and a detecting transducer composed of N periodic sections, the transfer admittance can be expressed by Eq. (6), where a_n, b_n=±1 denote the polarity of the electrode pair. Transducer configurations of a normal type and a phase-coded type will be discussed with a view of applying the surface wave transducer to a digital delay line. Fig. 8, 10 and 13 show the experimental impluse response of transducers using the normal sequence, Baker sequence and Golay sequence. With respect to the impulse response, a transducer that is phase-coded may be applicable to digital devices because the amplitude ratio of the main peak to the second sidelobed peak is large. Fig. 14 shows the experimental response of these transducers for random input pulses. Fig. 15 shows the eye-patterns which are obtained from a superposition of each impulse response corresponding to the random input pulses shown is Fig. 14 . From Fig. 15, it is seen that the transducer using the Golay sequence is more applicable to digital devices. In order to obtain the design method of the surface wave transducer for actual digital devices, the relations between several external conditions, such as the width of the input pulse and the load admittance, and the output response should be considered by computing the Fourier transform of the transmission characteristics as expressed in Eq. (20). Fig. 16 shows the maximum amplitude and the variation of peak position of the main response as a function of the width of the input pulse. From this figure, it is seen that the maximum output amplitude and the minimum variation of peak position are obtained for t=T(T=L/2V). Fig. 17 shows the experimental and calculated output response patterns as a function of the load admittance obtained from Eq. (20). Fig. 18 shows the instantaneous power (e_0(t)^2/R_<out>)as a function of the load admittance. From these figures, it is seen that(w_0C_tR_<out>)^<-1> is nearly equal to the maximum instantaneous power of the main response. Analysis gives a convenient means of calculating the relation between the input signals and the output response patterns, thus concisely demonstratig how transducer configurations using phase-coding, such as the Golay code, are useful for digital devices.
  • 山水 秀一郎, 中鉢 憲賢, 菊池 喜充
    原稿種別: 本文
    1974 年 30 巻 10 号 p. 541-548
    発行日: 1974/10/01
    公開日: 2017/06/02
    ジャーナル フリー
    In this paper, the results of experiments on two novel interdigital surface wave transducers with divided electrodes are presented and their properties are discussed. These divided electrode transducers are composed of minor transducers whose electrodes are connected in series. These electrodes are formed by dividing either the finger-length or finger-number of a conventional interdigital transducer(IDT) into m segments. The electrode patterns for the two types of transducers are shown in Figs. 1(divided finger-length type) and 2 (divided finger-number type). The equivalent circuit for an IDT composed of n spaces of finger electrodes can be represented by a cascade connection of alternately polarized equivalent circuits for the longitudinal vibrator as shown in Fig. 3. The fundamental equation of electroacoustic transformation for the acoustic terminal 1-1' in Fig. 3 is given by Eq. 1. Each coefficient is derived according to transmission-line theory. The force factor, A_n, and the damped admittance, Y_<dn>, are difined Eqs. 3 and 4, respectively. A characteristic of the divided-electrode transducers is that the free impedance for the divided finger-length type increases by m^2 compared to that for the undivided type, where m is the number of divided segments. For use in tuning, the condition for minimizing the effective attenuation is given by Eq. 12. A design formula in the form of Eq. 15 can be derived by substituting Eq. 14 fodr a coplanar electrode capacitance into Eq. 12. The transducer of the divided finger-number type consists of an array of several small transducers which are separated by suitable distances so that the generated waves are in phase with each other at each of the small transducers. When a transducer of the conventional type having a number n of intervals for finger electrodes is divided into m small transducers, the free impedance Z_<nm> is given by Eq. 18. Therefore, the Z_<nm> is increased m^2 times that of the conventional type. Experimental first, the divided finger-length electrode pair (m=3, n=11) and the undivided finger-length electrode pair (m=1, n=11) were made by photo-etching on a PZT ceramic substrate in order to compare their characteristics(Fig. 6). The characteristics obtaind for these transducers (center frequency:5. 3MHz)are shown in Figs. 7 and 8. Next, the divided finger-number electrode(m=2, n=15) and the undivided finger-number electrode (m=1, n=15) were made on each side of the undivided finger-number electrode for sending transducer on a PZT ceramic substrate (Fig. 10). The characteristics obtained for these transducers (center frequency:14. 5MHz) are shown in Figs. 11 and 12. In conclusion, from a theoretical analysis and experiments, it is clear that the real part of the free impedance increaces and input-output voltage ratio is improved with little sacrifice in the frequency bandwidth by dividing the interdigital electrodes and connecting them in series.
  • 佐藤 弘明, 目黒 敏靖, 山之内 和彦, 柴山 乾夫
    原稿種別: 本文
    1974 年 30 巻 10 号 p. 549-556
    発行日: 1974/10/01
    公開日: 2017/06/02
    ジャーナル フリー

    As the piezoelectric substrate for elastic surface wave filters and delay lines, 131° rotated Y-cut X-propagating crystalline lithium niobate plates are widely used because of their superiority in electromechanical coupling to Rayleigh waves and low beam steering compared to other cuts. However, an unknown spurious signal generated on the substrate frequentry prevents successful experiments. For filters, this leads to such phenomena for which the attenuation in the stop band cannot be guaranteed to be sufficiently large. This paper deals mainly with the experimental suppression of the spurious component through RF pulse responses and the frequency characteristics of the rotated Y-cut plates cut at several angles near 131°. The following facts are apparent:(1)The spurious component corresponds to the slower of the two shear waves propagating along the X-axis in the semiinfinite LiNbO_3 plate. (2)The component greatry depends on the cutting angle θ and is minimum at θ=128. 86°, that is, this angle gives the cut for optimum suppression. Experiments were performed using the plate rotated θdegree from the Y-axis about the X-axis of the LiNbO_3 crystal as illustrated in Fig. 1. The specimens were obtained by cutting or rubbing down at intervals of about one degree from 123. 6° to 131. 88° and their cut angles were measured exactly by X-ray diffraction. On the surface of these specimens, uniform overlap electrodes, shown in Fig. 2 were fabricated by the photolithographic technique and on the lower part of these elecrodes, many grooves were cut in order to suppress reflection Fig. 4 shows the RF pulse response of the θ=130. 86° specimen measured by the apparatus as shown in Fig. 3. Fig. 4(a) shows the response of the Rayleigh wave component when the amplitude is maximum and the frequency is 39. 5 MHz. Fig. 4(b) shows the response of the spurious component for maximum amplitude under the condition that the Rayleigh wave component is suppressed by adhering viscoelastic tape onto the propagating path and the frequency is 40. 5 MHz. By comparing these responses, it is clear that the spurious wave propagates faster than the Rayleigh wave by about 2. 5%. Next, several frequency responses were observed using a frequency spectrum analyzer under three conditions:(a) for a free path, (b) for suppression of the Rayleigh wave component by adhering viscoelastic tape onto the path and (c) for plastic clay set on the transducer electrode beside the tape, as shown in Fig. 5. These results are shown in Figs. 6, 7 and 8. The peak values of these responses are arranged and plotted in Fig. 9. The triangles in Figs. 11 and 12 show the velocities and the effective electromechanical coupling factors, respectively, as measured by the electrodes shown in Fig. 10. In these figures, the curves marked "×" shows the theoretical values calculated using the elastic and piezoelectric constants taken from the work of Waner, Onoe and Coquin, and the cueves marked "・" show the values calculated using the constant measured by Nakagawa, Yamanouchi and Shibayama. The propagation velocity of the spurious wave obtained from the relation between the pitch of the electrodes and the measured center frequency in Figs. 6, 7 and 8 is about 4050m/sec and this agrees well with the result for the RF pulse reponse mentioned above. It is conclided, by comparing the theoretical values shown in Fig. 11 with this result, that the spurious wave corresponds to the slower of the two shear waves in the LiNbO_3 crystals. It is clear from Fig. 9 that the spurious component is minimum for a cut angle of 127. 86°for which the plane corresponds to (0, -1, 4)-plane with a Bragg diffraction angle of 32. 63° and can be suppressed by about -60 dB with respect to the Rayleigh component. On the other hand , non-beam-steering properties of the new cut crystal were confirmed by measurements using a laser probe, as is shown in Fig.

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  • 吉川 昭吉郎, 石原 藤夫
    原稿種別: 本文
    1974 年 30 巻 10 号 p. 557-562
    発行日: 1974/10/01
    公開日: 2017/06/02
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  • 米山 務, 西田 茂穂
    原稿種別: 本文
    1974 年 30 巻 10 号 p. 563-568
    発行日: 1974/10/01
    公開日: 2017/06/02
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  • 中鉢 憲賢
    原稿種別: 本文
    1974 年 30 巻 10 号 p. 569-573
    発行日: 1974/10/01
    公開日: 2017/06/02
    ジャーナル フリー
  • 田村 光男, 米沢 正智
    原稿種別: 本文
    1974 年 30 巻 10 号 p. 574-577
    発行日: 1974/10/01
    公開日: 2017/06/02
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  • 安田 力
    原稿種別: 本文
    1974 年 30 巻 10 号 p. 578-583
    発行日: 1974/10/01
    公開日: 2017/06/02
    ジャーナル フリー
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